Since the present invention relates to imaging sensors and light-to-frequency converters, it is useful at this point to briefly review the dynamic range and noise characteristics of CMOS image sensors, and the operation of a light-to-frequency converter circuit.
The dynamic range and noise characteristics of CMOS image sensors will now be discussed. Until relatively recently, charge coupled devices (CCDs) maintained a dominant position in the field of digital imaging sensors. However, recent advances in the design and fabrication of complementary metal oxide semiconductor (CMOS) chips has meant that CMOS imaging sensors are beginning to adopt a more dominant position in the low-cost imaging market.
One of the main advantages of CMOS imaging sensors is that they can be produced using standard fabrication procedures which are already widely used for producing CMOS chips for computer processors, memory chips, etc. In contrast, CCDs require optimized charge transfer efficiency, and thus specialized fabrication facilities. Consequently, CMOS imaging sensor fabrication is considerably less expensive than CCD fabrication. Furthermore, in contrast with CCDs, the signal processing and control circuitry for a CMOS imaging sensor can be integrated directly onto the CMOS chip. However, the functionality and size economy provided by the extra circuitry on CMOS imaging sensor chips comes at the cost of increased dark current.
Dynamic range is a measurement of an imaging sensors ability to capture detail across a range of lighting conditions (i.e., from dark shadows to bright lighting). More specifically, referring to equation 1, an imaging sensor's dynamic range (DR) may be defined as the ratio of the sensor's largest non-saturating current (imax) to the smallest photo-current (imin) detectable by the sensor.
                              D          ⁢                                          ⁢          R                =                  20          ⁢                                          ⁢                      log            10                    ⁢                                    i              max                                      i              min                                                          (        1        )            
Consequently, from equation 1 it can be seen that to increase the dynamic range of an imaging sensor it is necessary to increase imax and decrease imin. However, referring to equation 2, the maximum non-saturating input current (imax) of an imaging sensor is a function of the sensor's well capacity (Qsat), dark current (idc) and integration time tint.
                              i          max                =                                            q              ⁢                                                          ⁢                              Q                sat                                                    t              int                                -                      i            dc                                              (        2        )            
Similarly, referring to equation 3, the smallest photo-current (imin) detectable by an imaging sensor is a function of the sensor's dark current, read noise σr2 and integration time.
                              i          min                =                              q                          t              int                                ⁢                                                                      1                  q                                ⁢                                  i                  dc                                ⁢                                  t                  int                                            +                              σ                r                2                                                                        (        3        )            
From equation 2, it can be seen that the largest non-saturating input signal (imax) of an imaging sensor can be increased by increasing the speed of the system (i.e., decreasing the integration time). However, referring to equation 3, decreasing the integration time of the imaging sensor also has the effect of increasing imin. Thus, the approach of decreasing the integration time of an imaging sensor will produce a limited increase in an imaging sensor's dynamic range.
Another method of increasing an imaging sensor's dynamic range is to use larger photodiodes (i.e., with larger well capacity). However, this has the effect of increasing the imaging sensor's dark current idcand thus increasing imin.
Dark current is the leakage current generated at the integration node of a photo-detector in the absence of any optical signal. Dark current originates from thermally generated electron-hole pairs that produce junction and transistor leakages. Accordingly, dark current is a function of temperature and junction area.
Operation of a light-to-frequency converter circuit will now be discussed. A light-to-frequency (LTF) converter, as disclosed in U.S. Pat. No. 5,850,195 is a CMOS imaging sensor with a large dynamic range.
The LTF converter comprises a control circuit 4, at least one photodiode 6 and a current to digital signal converter 8. The control circuit 4 controls the sensitivity of the LTF converter in accordance with a number of user-controllable input signals S0, S1, S2 and S3. The current to digital signal converter 8 uses a switched-capacitor charge metering technique to convert the photo-current generated by the photodiode(s) 6 to a digital signal of a specific frequency. In order to perform this conversion process, the current to digital signal converter 8 employs a bias circuit 10, a diode multiplexer circuit 12, an amplifier circuit 14, a switched capacitor network 16, a comparator 18 and a monostable multivibrator circuit 19.
The bias circuit 10 receives a control signal from the control circuit 4 to control the sensitivity of the current to digital signal converter 8. The bias circuit 10 transmits a further control signal to the diode multiplexer circuit 12, which also receives the photo-current generated by the photodiode(s) 6. The switched-capacitor charge metering technique employed by the current to digital signal converter 8 is implemented by the amplifier circuit 14, capacitor network 16 and monostable multivibrator circuit 19.
Referring to FIG. 2, the diode multiplexer circuit 12 receives a photo-current from the photodiode 6 and employs a charge sensing amplifier circuit 20 to effectively isolate the remaining circuitry of the current to digital signal converter 8 from the large capacitance of the photodiode 6 (<100 pF). The charge sensing amplifier 20 comprises an operational amplifier 22 configured in a closed loop configuration with its non-inverting input coupled to ground and a feedback capacitor 24 connected to the inverting input.
Since the operational amplifier 22 has a high input impedance, virtually no current flows through it. Consequently, since the non-inverting input of the operational amplifier 22 is connected to ground, the inverting input becomes a virtual ground. The output of the operational amplifier 24 changes to ensure that the inverting input of the operational amplifier 24 remains at the same potential as the non-inverting input. In the process, a current flows through the feedback capacitor 22 which has the same magnitude (but opposite sign) to the photo-current generated by the photodiode) 6 (ipd).
Equation 4 shows the relationship between the output voltage from the charge sensing amplifier 20 and the photo-current generated by the photodiode 6.
                              V          out                =                              -                          i              pd                                ⁢                                    T              int                                      C              fb                                                          (        4        )            
From the above expression it can be seen that the output voltage (Vout) from the charge sensing amplifier 20 is independent of the photodiode's 6 capacitance.
Returning to FIG. 1, the diode multiplexer circuit 12 transmits a current signal corresponding with the photo-current generated by the photodiode 6 (and processed by the charge sensing amplifier) to the amplifier circuit 14. The amplifier circuit 14 comprises at least one operational amplifier that integrates the voltage (Vout) generated from the photocurrent (ipd). Since the closed loop gain bandwidth product of an operational amplifier is inversely related to the capacitive loading at its inputs, the significant reduction in capacitive loading provided by the diode multiplexer circuit 12 (and its charge sensing amplifier) permits the use of a lower frequency operational amplifier in the amplifier circuit 14.
Furthermore, the reduction in capacitive loading made possible by the charge sensing amplifier in the diode multiplexer circuit 12 permits the use of larger photodiodes in the event of there being more than one photodiode 6 available. However, referring to FIG. 3, the increased output of the integrating amplifier (22) will reach a maximum value and be periodically reset.
FIG. 4 shows a system used for resetting the integrating amplifier in the amplifier circuit 14. In this system, the output voltage from the amplifier circuit 14 (Vout2) is transmitted to the comparator 18 of the current to digital signal converter 8. In the comparator 18, the output voltage (Vout2) is compared against a reference voltage (Vref). If the output voltage (Vout2) exceeds the reference voltage (Vref), the comparator 18 transmits a control signal (Ctrl) to the monostable multivibrator circuit 19. In response to the received control signal (Ctrl), the monostable multivibrator circuit 19 emits a pulsed signal, with each pulse being used to generate a charge to discharge the integrating amplifier 22.
The charge transferred to the integrating amplifier 22 with the emission of each pulse by the monostable multivibrator circuit 19 results in an average current that is equivalent to the photodiode current Ipd. Consequently, the frequency of the control signal Ctrl is also proportional to the photodiode current Ipd (assuming that the integrating amplifier in the amplifier circuit 14 settles completely during the period of the control signal). The control signal Ctrl is also fed to a divide-by-two circuit 30 to form the output signal from the LTF converter. Once the output voltage from the integrating amplifier (Vout2) is reduced to below the reference voltage (Vref), the monostable multivibrator circuit 19 is disabled.
FIG. 5 shows the timing of the reference voltage (Vref), output voltage from the integrating amplifier (Vout2), the control signal (Ctrl) from the comparator 18 and the overall output signal (Fout) from the LTF converter. Since, in accordance with equation 4, the slope of the charge sensing amplifier's output is proportional to the incident light, the frequency of the output signal (Fout) from the light-to-frequency converter is also proportional to the incident light intensity. This proportionality is more clearly expressed in equation 5.
                              F          out                =                              i            pd                                2            ⁢                                          C                fb                            ⁡                              (                                                      V                    ref                                    -                                      V                    rt                                                  )                                                                        (        5        )            
The dynamic range of this system is Foutmax-Foutmin. The maximum frequency (Foutmax) is typically determined by the maximum operating speed of the integrating amplifier in the amplifier circuit 14, which sets the maximum slew rate (m in FIG. 5). The maximum frequency (Foutmax) can be relatively easily influenced by the design of the on-board operational amplifiers (e.g., by reducing the parasitic capacitance of the operational amplifiers or increasing the bias currents of the operational amplifiers). However, the minimum frequency (Foutmin) is typically determined by the dark current flowing through the photodiode 6.
FIGS. 6 and 7 show top plan views of conventional LTF converter circuits comprising an LTF conversion circuit section 32 and a photo-generated electron collection section 33. The LTF conversion circuitry 32 (e.g., control circuit and current to digital signal converter) comprises NMOS 34 and PMOS 36 transistors embedded in a P-well 38 and N-well 40, respectively. In FIG. 6 the photo-generated electron collection section 33 comprises a large N-well photodiode 42 surrounding an N+ contact 43. In FIG. 7, the photo-generated electron collection section 33 comprises a large N+ photodiode 44. The photodiodes 42 or 44 are to be isolated from the LTF conversion circuitry 32 by a P-well material 46. Finally, the photo-current collected from the photodiode 42 or 44 is transmitted to the LTF conversion circuitry section 32 by a metal contact 48.
These above traditional LTF converter designs typically have good light sensitivity. However, referring to FIG. 8, these devices also suffer from a large dark current because of the large photodiode junction (i.e., the boundary between the N+ 44 photodiode or N-well 42 photodiode and the P-well 46). In particular, it will be noted that the area of the photo-generated electron collection section 33 (photodiode 42/44) is equal to the area of the LTF conversion circuitry section 32. In addition, the photo-generated electron collection section 33 (photodiode 42/44) also possesses a large perimeter. These two attributes increase the dark current of a conventional LTF converter, and thereby limit the low-light performance of the device.
From the above it can be seen that while increasing the size of the photodiodes in an LTF converter increases the imaging sensor's sensitivity, it also decreases the sensor's dynamic range by increasing the photodiode's dark current.